Integrated circuit employable with a power converter

ABSTRACT

An integrated circuit employable with a power converter. In one embodiment, the integrated circuit includes a power switch of a power train of the power converter formed on a semiconductor substrate. The integrated circuit also includes a driver switch of a driver configured to provide a drive signal to the power switch and embodied in a transistor including a gate located over a channel region recessed into the semiconductor substrate. The transistor also includes a source/drain including a lightly doped region located adjacent the channel region and a heavily doped region located adjacent the lightly doped region. The transistor further includes an oppositely doped well located under and within the channel region. The transistor still further includes a doped region, located between the heavily doped region and the oppositely doped well, having a doping concentration profile less than a doping concentration profile of the heavily doped region.

TECHNICAL FIELD

The present invention is directed, in general, to integrated circuitsand, more specifically, to an integrated circuit incorporating highervoltage devices and low voltage devices therein.

BACKGROUND

The design of early integrated circuits focused on implementing anincreasing number of small semiconductor devices on a semiconductorsubstrate to achieve substantial improvements in manufacturingefficiency and cost, product size, and performance. The continuingimprovements in the design of integrated circuits over the past fewdecades has been so dramatic and so pervasive in numerous products thatthe effects can be measured in changes in industries.

The design and construction of integrated circuits has continued toevolve in a number of different areas. One area of innovation is acontinuing reduction of feature sizes of semiconductor devices such ascontrol and signal processing devices formed on a semiconductorsubstrate. Another area of innovation is the advent of constructiontechniques to incorporate higher voltage semiconductor devices (alsoreferred to as “higher voltage devices”) having higher voltage handlingcapability such as switches of a power train of a power converter intothe integrated circuits.

An objective of incorporating control and signal processing devices on asemiconductor substrate with the higher voltage devices often encountersconflicting design requirements. More specifically, lower voltages(e.g., 2.5 volts) are employed with the control and signal processingdevices (hence, also referred to as “low voltage devices”) to preventflashover between the fine line structures thereof. A potentialdifference of only a few volts separated by a fraction of a micrometercan produce electric fields of sufficient magnitude to induce locallydestructive ionization in the control and signal processing devices.

When employing the higher voltage devices therewith, it is oftennecessary to sense and switch higher external circuit voltages (e.g.,ten volts or higher) on the integrated circuit. To accommodate thehigher voltage devices on a semiconductor substrate with the control andsignal processing devices, a large number of processing steps areperformed to produce the integrated circuit. Since the cost of anintegrated circuit is roughly proportional to the number of processingsteps to construct the same, there has been limited progress in theintroduction of low cost integrated circuits that include both controland signal processing devices and higher voltage devices such as theswitches of the power train of a power converter.

The aforementioned constraints have been exacerbated by the need toemploy a substantial area of the semiconductor substrate to incorporatemore efficient and even higher voltage devices into an integratedcircuit. Inasmuch as the cost of a die that incorporates the integratedcircuit is roughly proportional to the area thereof, the presence of thehigher voltage devices conflicts with the reduction in area achieved byincorporating the fine line features in the control and signalprocessing devices.

With respect to the type of semiconductor devices readily available,complementary metal oxide semiconductor (“CMOS”) devices are commonlyused in integrated circuits. The CMOS devices such P-type metal oxidesemiconductor (“PMOS”) devices and N-type metal oxide semiconductor(“NMOS”) devices are used as logic devices, memory devices, or otherdevices such as the control and signal processing devices. In additionto the CMOS devices, laterally diffused metal oxide semiconductor(“LDMOS”) devices such as P-type laterally diffused metal oxidesemiconductor (“P-LDMOS”) devices and N-type laterally diffused metaloxide semiconductor (“N-LDMOS”) devices are also commonly used inintegrated circuits. LDMOS devices are generally used for the highervoltage devices in the integrated circuit. In the context of CMOStechnology, the higher voltage devices generally relate to devices thatoperate at voltages above a standard operating voltage for the selectedCMOS devices (e.g., the low voltage devices). For instance, CMOS devicesemploying fine line structures having 0.25 micrometer line widthsoperate at or below about 2.5 volts. Thus, higher voltage devicesgenerally include any devices operating above approximately 2.5 volts.

Integrating the CMOS and LDMOS devices on a semiconductor substrate hasbeen a continuing goal in the field of microelectronics and has been thesubject of many references over the years. For instance, U.S. Pat. No.6,541,819 entitled “Semiconductor Device Having Non-Power Enhanced andPower Enhanced Metal Oxide Semiconductor Devices and a Method ofManufacture Therefor,” to Lotfi, et al., issued Apr. 1, 2003, which isincorporated herein by reference, incorporates non-power enhanced metaloxide semiconductor devices (i.e., low voltage devices) with powerenhanced metal oxide semiconductor devices (i.e., higher voltagedevices) on a semiconductor substrate. While Lotfi, et al. provides aviable alternative to integrating low voltage devices and higher voltagedevices on the semiconductor substrate, further improvements arepreferable in view of the higher voltage handling capability associatedwith the use of higher voltage devices such as with the LDMOS devices inthe power train of a power converter.

In the field of power microelectronics, the CMOS devices may be employedas the control and signal processing devices integral to the controllerof a power converter. As an example, the control and signal processingdevices are employed as low voltage switches and comparators that formportions of the controller of the power converter. The LDMOS devices, onthe other hand, may be employed as the higher voltage devices integralto the power train of the power converter. The higher voltage devicesperform the power switching functions to control the flow of power to,for instance, a microprocessor. The power switches include the mainpower switches, synchronous rectifiers, and other power switches germaneto the power train of the power converter. The power switches can alsobe used for circuit protection functions such as a rapidly actingelectronic version of an ordinary fuse or circuit breaker. Variations ofpower switches include metal oxide semiconductor field effecttransistors (“MOSFETs”) that exhibit low level gate-to-source voltagelimits (e.g. 2.5 volts) and otherwise are capable of handing the highervoltages germane to the power train of the power converter.

To achieve the overall reduction in size, the integrated circuits asdescribed herein should include control and signal processing deviceswith fine line structures having sub micron line widths (e.g., 0.25micrometers) on a semiconductor substrate that operate with lowervoltages to prevent flashover within the integrated circuit. At the sametime, the integrated circuit may incorporate higher voltage devices thatcan conduct amperes of current and withstand voltages of, for instance,ten volts. A benefit of incorporating the low voltage devices and thehigher voltage devices on the semiconductor substrate is that it ispossible to accommodate higher switching frequencies in the design ofthe power processing circuit due to a reduction of parasiticcapacitances and inductances in the integrated circuit.

While a design and implementation of low voltage devices such as logicdevices that form portions of a microprocessor have been readilyincorporated into integrated circuits, the systems that power the logicdevices have not to date been readily incorporated into integratedcircuits. There has been pressure directed to the power electronicsindustry to make parallel improvements in the power conversiontechnology and, in particular, with the power converters that regulatethe power to, for instance, the microprocessors that employ a high levelof integrated circuit technology in the design thereof. Thus, anevolutionary direction in the power electronics industry is to reducethe size and cost of the power converters which correspondingly inducesgreater levels of silicon integration in a design of the integratedcircuits embodying the same.

Although power converters have shown dramatic improvements in size,cost, and efficiency over the past few decades, the design of the powerconverters have not kept pace with the improvements in integratedcircuit technology directed to the logic devices and the like, whichfollow Moore's Law demonstrating a doubling of performance every 18months as viewed by certain metrics of digital performance. Asrepresentative examples of improvements in the smaller and more compactpower converters, see U.S. Pat. No. 5,469,334, entitled “PlasticQuad-packaged Switched-mode Integrated Circuit with IntegratedTransformer Windings and Mouldings for Transformer Core Pieces,” toBalakrishnan, issued on Nov. 21, 1995, and U.S. Pat. No. 5,285,369,entitled “Switched Mode Power Supply Integrated Circuit with Start-upSelf-biasing,” to Balakrishnan, issued on Feb. 8, 1994, which areincorporated herein by reference. While Balakrishnan and otherreferences have demonstrated noticeable improvements of incorporatingpower converters into an integrated circuit, an industry wideintegration of higher voltage level devices (again, such as the switchesof the power train) into the design of integrated circuits, especiallyin power converters, has not yet gained industry wide adoption.

Another issue in a design of the power converters is an increase of theswitching frequency (e.g., five megahertz) of the power train thereof.The energy stored in reactive circuit elements (e.g., inductors andcapacitors) associated with the power converter is inverselyproportional to the switching frequency, and the size of the reactivecircuit elements is also correspondingly inversely proportional to theswitching frequency. A power converter is generally designed to handlethe highest switching frequency without significantly compromising powerconversion efficiency. Otherwise, the switching frequency could besimply increased with a consequent reduction in the size and cost of thepower converter. Achieving a high switching frequency is dependent onreducing the parasitic circuit elements such as stray interconnectioncapacitance and inductance. As mentioned above, incorporating the lowvoltage devices and the higher voltage devices within an integratedcircuit embodying the power converter can have a significant impact inreducing the interconnection paths and consequently the strayinterconnection parasitic capacitance and inductance. Additionally,reducing the inherent parasitic losses in the switches of the powerconverter such as energy stored in a gate of a MOSFET can also have asignificant impact on the switching frequency of the power converter.

Accordingly, what is needed in the art is an integrated circuit andmethod of forming the same that incorporates higher voltage devices andlow voltage devices on a semiconductor substrate that overcomes thedeficiencies in the prior art. Additionally, there is a need in the artfor a higher voltage device (e.g., a transistor such as a LDMOS device)that can accommodate higher voltages and is capable of being integratedwith low voltage devices on a semiconductor substrate in an integratedcircuit that may form a power converter or portions thereof.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, andtechnical advantages are generally achieved, by advantageous embodimentsof the present invention which, includes an integrated circuit formed ona semiconductor substrate and employable with a power converter. In oneembodiment, the integrated circuit includes a power switch of a powertrain of the power converter formed on a semiconductor substrate. Theintegrated circuit also includes a complementary metal oxidesemiconductor device employable in a controller configured to provide asignal to control a duty cycle of the power switch and formed on thesemiconductor substrate. The integrated circuit also includes a driverswitch of a driver configured to provide a drive signal to the powerswitch as a function of the signal from the controller and embodied in atransistor including a gate located over a channel region recessed intothe semiconductor substrate. The transistor also includes a source/drainincluding a lightly doped region located adjacent the channel region anda heavily doped region located adjacent the lightly doped region. Thetransistor further includes an oppositely doped well located under andwithin the channel region. The transistor still further includes a dopedregion, located between the heavily doped region and the oppositelydoped well, having a doping concentration profile less than a dopingconcentration profile of the heavily doped region.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated by those skilled in the art that the conceptionand specific embodiment disclosed may be readily utilized as a basis formodifying or designing other structures or processes for carrying outthe same purposes of the present invention. It should also be realizedby those skilled in the art that such equivalent constructions do notdepart from the spirit and scope of the invention as set forth in theappended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, reference isnow made to the following descriptions taken in conjunction with theaccompanying drawings, in which:

FIG. 1 illustrates a diagram of an embodiment of a power converterembodied in, or portions thereof, an integrated circuit constructedaccording to the principles of the present invention;

FIG. 2 illustrates a schematic diagram of an embodiment of a controllerin an environment of a power converter embodied in, or portions thereof,an integrated circuit constructed according to the principles of thepresent invention;

FIG. 3 illustrates a schematic diagram of an embodiment of a driver of apower converter embodied in, or portions thereof, an integrated circuitconstructed according to the principles of the present invention;

FIG. 4 illustrates a cross sectional view of an embodiment of asemiconductor device employable in an integrated circuit constructedaccording to the principles of the present invention;

FIGS. 5A to 5H illustrate cross sectional views of an embodiment ofconstructing a micromagnetic device employable in an integrated circuitconstructed according to the principles of the present invention;

FIG. 6 illustrates an isometric view of an embodiment of a micromagneticdevice employable in an integrated circuit constructed according to theprinciples of the present invention; and

FIG. 7 illustrates a cross sectional view of an embodiment of an outputfilter employable in an integrated circuit constructed according to theprinciples of the present invention.

Corresponding numerals and symbols in the different figures generallyrefer to corresponding parts unless otherwise indicated. The figures aredrawn to clearly illustrate the relevant aspects of the preferredembodiments and are not necessarily drawn to scale.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the presently preferred embodiments arediscussed in detail below. It should be appreciated, however, that thepresent invention provides many applicable inventive concepts that canbe embodied in a wide variety of specific contexts. The specificembodiments discussed are merely illustrative of specific ways to makeand use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferredembodiments in a specific context, namely, an integrated circuitincluding a transistor [e.g., embodied in a laterally diffused metaloxide semiconductor (“LDMOS”) device] and methods of forming the same.While the principles of the present invention will be described in theenvironment of a power converter, any application that may benefit froma transistor that can accommodate higher voltages and is integrable witha low voltage device [e.g., complementary metal oxide semiconductor(“CMOS”) device] on a semiconductor substrate is well within the broadscope of the present invention.

The advantages associated with incorporating the higher voltage LDMOSdevices with the low voltage CMOS devices facilitate the ongoingincorporation of integrated circuits with higher levels of integrationinto more products such as power converters. For the purposes of thepresent invention, higher voltage devices refer to devices that canaccommodate higher operating voltages than the standard operatingvoltages for a referenced low voltage device. As an example and in thecontext of CMOS technology, the higher voltage devices generally relateto devices that operate at voltages above a standard operating voltagefor the selected CMOS devices (e.g., the low voltage devices). Forinstance, CMOS devices employing fine line structures having 0.25micrometer line widths operate at or below about 2.5 volts. Thus, highervoltage devices generally include any devices operating aboveapproximately 2.5 volts. In yet another context, the higher voltagedevices also generally include devices that may exhibit a low levelgate-to-source voltage limit (e.g., 2.5 volts) and, at the same time,can handle drain-to-source voltages above the gate-to-source voltagelimit thereof (e.g., ten volts).

Referring initially to FIG. 1, illustrated is a diagram of an embodimentof a power converter including a semiconductor device constructedaccording to the principles of the present invention. The powerconverter includes a power train 110, a controller 120 and a driver 130,and provides power to a system such as a microprocessor. While in theillustrated embodiment, the power train 110 employs a buck convertertopology, those skilled in the art should understand that otherconverter topologies such as a forward converter topology are wellwithin the broad scope of the present invention.

The power train 110 of the power converter receives an input voltageV_(in) from a source of electrical power (represented by a battery) atan input thereof and provides a regulated output voltage V_(out) topower, for instance, a microprocessor at an output of the powerconverter. In keeping with the principles of a buck converter topology,the output voltage V_(out) is generally less than the input voltageV_(in) such that a switching operation of the power converter canregulate the output voltage V_(out). A main switch Q_(mn) [e.g., aP-channel metal oxide semiconductor field effect transistor (“MOSFET”)embodied in a P-type laterally diffused metal oxide semiconductor(“P-LDMOS”) device] is enabled to conduct for a primary interval(generally co-existent with a primary duty cycle “D” of the main switchQ_(mn)) and couples the input voltage V_(in) to an output filterinductor L_(out). During the primary interval, an inductor currentI_(Lout) flowing through the output filter inductor L_(out) increases asa current flows from the input to the output of the power train 110. AnAC component of the inductor current I_(Lout) is filtered by the outputcapacitor C_(out).

During a complementary interval (generally co-existent with acomplementary duty cycle “1-D” of the main switch Q_(mn)), the mainswitch Q_(mn) is transitioned to a non-conducting state and an auxiliaryswitch Q_(aux) [e.g., a N-channel MOSFET embodied in a N-type laterallydiffused metal oxide semiconductor (“N-LDMOS”) device] is enabled toconduct. The auxiliary switch Q_(aux) provides a path to maintain acontinuity of the inductor current I_(Lout) flowing through the outputfilter inductor L_(out). During the complementary interval, the inductorcurrent I_(Lout) through the output filter inductor L_(out) decreases.In general, the duty cycle of the main and auxiliary switches Q_(mn),Q_(aux) may be adjusted to maintain a regulation of the output voltageV_(out) of the power converter. Those skilled in the art shouldunderstand, however, that the conduction periods for the main andauxiliary switches Q_(mn), Q_(aux) may be separated by a small timeinterval to avoid cross conduction therebetween and beneficially toreduce the switching losses associated with the power converter.

The controller 120 of the power converter receives a desiredcharacteristic such as a desired system voltage V_(system) from aninternal or external source associated with the microprocessor, and theoutput voltage V_(out) of the power converter. The controller 120 isalso coupled to the input voltage V_(in) of the power converter and areturn lead of the source of electrical power (again, represented by abattery) to provide a ground connection therefor. While only a singleground connection is illustrated in the present embodiment, thoseskilled in the art should understand that multiple ground connectionsmay be employed for use within the controller 120. A decouplingcapacitor C_(doc) is coupled to the path from the input voltage V_(in)to the controller 120. The decoupling capacitor C_(dec) is configured toabsorb high frequency noise signals associated with the source ofelectrical power to protect the controller 120.

In accordance with the aforementioned characteristics, the controller120 provides a signal (e.g., a pulse width modulated signal S_(PWM)) tocontrol a duty cycle and a frequency of the main and auxiliary switchesQ_(mn), Q_(aux) of the power train 110 to regulate the output voltageV_(out) thereof. The controller 120 may also provide a complement of thesignal (e.g., a complementary pulse width modulated signal S_(1-PWM)) inaccordance with the aforementioned characteristics. Any controlleradapted to control at least one switch of the power converter is wellwithin the broad scope of the present invention. As an example, acontroller employing digital circuitry is disclosed in U.S. patentapplication Ser. No. 10/767,937, entitled “Controller for a PowerConverter and a Method of Controlling a Switch Thereof,” to Dwarakanath,et al. and U.S. patent application Ser. No. 10/766,983, entitled“Controller for a Power Converter and Method of Controlling a SwitchThereof,” to Dwarakanath, et al., which are incorporated herein byreference.

The power converter also includes the driver 130 configured to providedrive signals S_(DRV1), S_(DRV2) to the main and auxiliary switchesQ_(mn), Q_(aux), respectively, based on the signals S_(PWM), S_(1-PWM)provided by the controller 120. There are a number of viablealternatives to implement a driver 130 that include techniques toprovide sufficient signal delays to prevent crosscurrents whencontrolling multiple switches in the power converter. The driver 130typically includes switching circuitry incorporating a plurality ofdriver switches that cooperate to provide the drive signals S_(DRV1),S_(DRV2) to the main and auxiliary switches Q_(mn), Q_(aux). Of course,any driver 130 capable of providing the drive signals S_(DRV1), S_(DRV2)to control a switch is well within the broad scope of the presentinvention.

According to the principles of the present invention, the main andauxiliary switches Q_(mn), Q_(aux) are power switches that can beincorporated into a semiconductor device in an integrated circuitproximate control or signal processing devices that perform many of thecontrol functions of the controller 120 of the power converter. Asmentioned above, the control and signal processing devices are typicallyCMOS devices such as P-type metal oxide semiconductor (“PMOS”) devicesand N-type metal oxide semiconductor (“NMOS”) devices. The PMOS and NMOSdevices may also be referred to as P-channel and N-channel MOSFETs,respectively. Lower voltages (e.g., 2.5 volts) are employed with thecontrol and signal processing devices (hence, also referred to as “lowvoltage devices”) to prevent flashover between the fine line structuresthereof.

The main and auxiliary switches Q_(mn), Q_(aux) of the power train 110,ones of the plurality of driver switches of the driver 130 and selectedswitches or other devices within the controller 120 are typically formedby LDMOS devices that handle higher voltages (e.g., ten volts) and henceare referred to as higher voltage devices. Integrating the control andsignal processing devices, power switches and other switches (e.g., thedriver switches) on a semiconductor substrate provides opportunities forsubstantial reductions in cost and size of an integrated circuitemployable with the power converter or other apparatus employing likedevices.

Additionally, when providing a drive signals S_(DRV1), S_(DRV2) to aswitch (e.g., the main switch Q_(mn)) such as a P-channel MOSFET havinga control voltage limit (i.e., a gate voltage limit) of 2.5 volts, andin the environment of a power converter having a nominal input voltageV_(in) of five volts, the extended voltage range present on the gateterminal of the main switch Q_(mn) may break down the integrity of thethin gate oxide thereof. In other words, when the input voltage V_(in)to the power converter which is translated into the drive signalS_(DRV1) to the main switch Q_(mn) under certain conditions as describedabove exceeds the gate voltage limit thereof, the main switch Q_(mn) maybe damaged and fail. Another layer of complexity arises when theplurality of driver switches of the driver 130 are referenced to avoltage level (e.g., a ground potential) and the main switch Q_(mn) tobe driven is referenced to another voltage (e.g., the input voltageV_(in) to the power converter). Colloquially, the main switch Q_(mn) ofthe power converter is referred to as a “floating” switch. A driver 130for the power converter, therefore, should be capable of handlingapplications wherein the main switch Q_(mn) to be driven exhibits asmaller control voltage limit (e.g., gate voltage limit) from thecontrol terminal to another terminal (e.g., the gate terminal to thesource terminal) thereof and is referenced to a voltage level differentfrom the driver 130.

Turning now FIG. 2, illustrated is a schematic diagram of an embodimentof a controller in an environment of a power converter embodied in, orportions thereof, an integrated circuit constructed according to theprinciples of the present invention. The power converter includes acontroller 210, a driver 220 and a power train 230. The controller 210provides a signal (e.g., a pulse width modulated signal S_(PWM)) tocontrol a duty cycle and a frequency of main and auxiliary switchesQ_(mn), Q_(aux) of the power train 230 to regulate an outputcharacteristic (e.g., an output voltage V_(out)) thereof. The controller210 may also provide a complement of the signal (e.g., a complementarypulse width modulated signal S_(1-PWM)) via a level shift and crossovercircuit 237. The level shift and crossover control circuit 237 is alsoadapted to adjust a delay between the signals S_(PWM), S_(1-PWM) thatcontrol the duty cycle of the main and auxiliary switches Q_(mn),Q_(aux) to substantially prevent a cross conduction and enhance theswitching transitions therebetween.

The power train 230 employs a buck converter topology, which has beendescribed above with respect to FIG. 1. The driver (e.g., a levelshifting gate driver) 220 provides gate drive signals S_(DRV1), S_(DRV2)for the main and auxiliary switches Q_(mn), Q_(aux), and also for asense switch (also referred to as a “switch in a controller 210 of thepower converter,” e.g., a P-channel MOSFET embodied in a P-LDMOS device,or also referred to as “another LDMOS device”) Q_(s). The sense switchQ_(s) is configured to measure an output characteristic (e.g., an outputcurrent) of the power converter.

The low voltage and higher voltage devices of the power converter may beembodied in a semiconductor device as illustrated and described withrespect to FIG. 4 to form portions of a power converter embodied in anintegrated circuit. Additionally, ones of the devices of the powerconverter such as an output inductor L_(out) and output capacitorC_(out) of the power train 230, and a soft-start capacitor C_(ss) and aselect resistor R_(select) (which selects a set point for the outputvoltage V_(out)) associated with the controller 210 may be discretedevices incorporated into or with an integrated package with thesemiconductor devices that embody other devices of the power converterand still remain within the broad scope of the present invention. Thediscrete devices are often employed with the power converter to provideapplication flexibility to allow for cost reductions and design optionsin constructing the power converter.

The controller 210 is coupled to the input voltage V_(in) and the outputvoltage V_(out) of the power converter and to first and second groundconnections PGND, AGND. For a representative power converter, the inputvoltage V_(in) is unregulated and falls within an operational range of2.5 to 6.5 volts. The output voltage V_(out) is well regulated (e.g.,within a three percent tolerance) and can be adjusted between, forinstance, 1.2 to 3.5 volts. The controller 210 of the power converteralso receives a desired characteristic such as a desired system voltageV_(system) from an internal or external source associated with, forinstance, a microprocessor powered by the power converter.

A soft start operation of the power converter may be adjusted by aselection of a soft start capacitor C_(ss), and the output voltageV_(out) may be adjusted by the select resistor R_(select). A signalindicating a normal operation of the power converter is provided via apower good connection PWRGD. The active devices of the power converterare powered from the input voltage V_(in) or from an internal, regulatedvoltage source, configured as a linear regulator 235 coupled to theinput voltage V_(in). The linear regulator 235 can be implemented as adissipative regulator as hereinafter described.

As is well understood by those skilled in the art, the first and secondground connections PGND, AGND are representative of ground connectionsfor the higher voltage devices handling higher currents and the lowvoltage devices handling low currents, respectively. The first groundconnection PGND is for currents flowing in the higher voltage devicesthat are less sensitive to system noise. The second ground connectionAGND is for currents flowing in the low voltage devices that are moresensitive to system noise. The first and second ground connections PGND,AGND are typically coupled at a single point within the power converter.

As described herein and, more specifically, with respect to FIG. 4below, the low voltage devices are generally embodied in PMOS and NMOSdevices which may be integrable with the higher voltage devices embodiedin P-LDMOS and N-LDMOS devices in a semiconductor device. As a result,the power converter is more readily incorporated into an integratedcircuit. Additionally, bias voltages V_(bias) (which may be internallyor externally generated) are resident throughout the controller 210. Thehigher voltage devices within the power converter operate from a highervoltage source such as the input voltage V_(in), and the low voltagedevices operate from a low voltage source which is usually wellregulated such as the bias voltages V_(bias). The voltage sourceconnections within the power converter are not intended to beexhaustive, but rather indicative of possible operational voltages forthe particular devices of the power converter.

An exemplary operation of the controller 210 will hereinafter bedescribed. A switching frequency of the power train 230 is generated bya sawtooth generator 240, which may be implemented using a currentsource to charge a capacitor coupled to a comparator (not shown). Whenthe voltage of the capacitor exceeds a threshold value, the comparatorenables a switch (not shown), quickly discharging the capacitor. Thecharge and discharge process regularly repeats, generating a sawtoothwaveform for the voltage across the capacitor. To provide a consistentswitching frequency, the sawtooth generator 240 is generally poweredfrom an internal, regulated voltage source providing the bias voltageV_(bias). A trim resistor R_(trim) may be included to adjust theswitching frequency during the design and manufacture of the controller210. For a better understanding of sawtooth generators, see “The Art ofElectronics,” by Horowitz, et al., Cambridge University Press, SecondEdition, pp. 288-291, 1989, the entire reference being incorporatedherein by reference.

The output voltage V_(out) is coupled through a compensation network 245to a non-inverting input of an error amplifier 250 of the controller210. Alternatively, the voltage representing the output voltage V_(out)may be determined from a remote location in a distribution network andprovided to the error amplifier 250. The error amplifier 250 is furthercompensated by a feedback network represented by a compensationcapacitor C_(comp). More extensive compensation networks can be providedas necessary for the error amplifier 250 as the application dictates.The compensation network 245 is coupled, via a select connection SEL, tothe select resistor R_(select), which is coupled to the second groundconnection AGND. The select resistor R_(select) provides an option toselect the set point for the output voltage V_(out) for the powerconverter.

The output of the error amplifier 250 is coupled to the non-invertinginput of a comparator (e.g., a PWM comparator) 255 that compares theoutput of the error amplifier 250 with an output of the sawtoothgenerator 240. An output of the PWM comparator 255 is high during aprimary interval when the main switch Q_(mn) of the power train isconfigured to conduct. The output of the PWM comparator 255 is lowduring a complementary interval when the main switch Q_(mn) of the powertrain is transitioned to a non-conducting state and the auxiliary switchQaux is configured to conduct. A non-inverting input of the erroramplifier 250 is coupled to a bandgap reference circuit 260 thatsupplies a well-regulated voltage (e.g., 1.07 volts) and a referencevoltage selector 265. The reference voltage selector 265 provides areference voltage V_(ref) to the non-inverting input of the erroramplifier 250 to establish a reference comparison for regulating theoutput voltage V_(out) of the power converter.

The bandgap reference circuit 260 preferably uses bipolar CMOStechnology and includes a disable pin (not shown) to disable an outputtherefrom. For example, when the disable pin is pulled high, the outputfrom the bandgap reference circuit 260 can be pulled close to a groundpotential with a switch (not shown), thereby disabling an operation ofthe power converter. The compensation network 245, as indicated above,is coupled to the select resistor R_(select) to provide the set pointfor the output voltage V_(out). The select resistor R_(select) may becoupled and operative in parallel with a resistor in a voltage dividernetwork 247 to control a fraction of the output voltage V_(out) therebyfurther refining a set point for the output voltage V_(out) for thepower converter. The use of voltage dividers, in general, to alter setpoints is well understood in the art and will not herein be described.

A soft start operation of the power converter is controlled, in part, bya soft start capacitor C_(ss). During a start up period of the powerconverter, the output voltage V_(out) of the power converter issubstantially zero, whereas during normal operation, a control loop ofthe controller 210 controls the conduction periods of the main andauxiliary switches Q_(mn), Q_(aux) to provide a regulated output voltageV_(out). When the main switch Q_(mn) is initially enabled to conduct andthe auxiliary switch Q_(aux) is non-conducting, a substantial in-rushcurrent to the power converter may occur in accordance with the inputvoltage V_(in) to charge the output capacitor C_(out). This conditionmay produce a substantial overshoot of the output voltage V_(out) as theoutput inductor L_(out) and output capacitor C_(out) resonantly ring inresponse to the in-rush current.

Thus, a slowly increasing set point for the control loop during thestart up period is preferable and can be achieved by increasing avoltage across the soft start capacitor C_(ss) at a controlled rate.During an initial operation of the power converter (and/or during are-start operation), the soft start capacitor C_(ss) is charged by acurrent source 270 (via a soft start connection SS), which is coupled tothe reference voltage selector 265. The reference voltage selector 265compares a voltage across the soft start capacitor C_(ss) with a voltageprovided by the bandgap reference circuit 260 and the system voltageV_(system) and selects the smaller value therefrom. The resultingreference voltage V_(ref) from the reference voltage selector 265 isprovided to the non-inverting input of the error amplifier 250 toregulate the set point for the output voltage V_(out) for the powerconverter.

Thus, during the soft start operation and when the voltage across thesoft start capacitor C_(ss) is smaller than the voltage of the bandgapreference circuit 260, the voltage across the soft start capacitorC_(ss) controls and slowly ramps up according to the charging rate ofthe soft start capacitor C_(ss). When the voltage across the soft startcapacitor C_(ss) exceeds the voltage from the bandgap reference circuit260, the voltage from the bandgap reference circuit 260 provides thecontrolling signal for the reference voltage V_(ref) to the erroramplifier 250. As an example, the value of the soft start capacitorC_(ss) is 15 nanofarads and the current source 270 provides about 10microamperes of current. This combination results in a rate of increaseof the voltage across the soft start capacitor of about 0.67volts/millisecond. Inasmuch as an inverting input to a soft startcomparator 275 is about, for instance, 0.8 volts, a time delay of about1.2 milliseconds is sustained before a soft start AND gate 280 enables aswitching operation of the power train 230 of the power converter. Ofcourse, the period of delay can be altered by changing the value of thesoft start capacitor C_(ss) or the value of the current source 270.

The linear regulator 235 provides a well regulated, low voltage biasvoltage V_(bias) (e.g., 2.5 volts) to supply power for the low voltagedevices having voltage limitations as generally determined by fine linesemiconductor structures thereof. The linear regulator 235 is poweredfrom the input voltage V_(in) and is coupled to a bypass capacitorC_(bp). The bypass capacitor C_(bp) can be formed from a semiconductordevice as described herein or by other device techniques and structures.Additional bypass capacitors C_(bp) may be employed within thecontroller 210 and power converter, in general, to absorb system noisetherein.

The linear regulator 235 preferably includes a higher voltage deviceimplemented with an N-LDMOS device acting as a series-pass, regulatingswitch (not shown). An operational amplifier (not shown) is included inthe linear regulator 235 that senses the bias voltage V_(bias) and areference voltage such as provided by the bandgap reference circuit 260to provide negative feedback to a control terminal of the series-pass,regulating switch, thereby providing voltage regulation for the biasvoltage V_(bias). The design of dissipative linear regulators 235 with afeedback control are well known in the art and will not herein bedescribed. For a better understanding of the design of dissipativelinear regulators, see chapter six of Horowitz, et al.

A number of circuits such as protection circuits within the controller210 disable an operation of the power converter during unusual orundesirable operating conditions. The output of the circuits arecombined via AND logic gates with an output from the PWM comparator 255to disable the operation of the power converter when necessary. Athermal shutdown circuit 282 monitors a temperature of the powerconverter (e.g., a temperature of the switch embodied in a semiconductordevice located on a semiconductor substrate) to protect, for example,against a possible low impedance circuit coupled inadvertently across anoutput of the power train 230. The temperature monitoring function canbe provided using a voltage reference (not shown) which is dependent,preferably linearly, on the temperature of the monitored portion of thepower converter. An output of the voltage reference is compared, forinstance, with the output of the bandgap reference circuit 260 using acomparator (not shown) and, when there is a sufficient voltagedifference therebetween, the comparator switches and provides a signalto a protection circuit AND gate 284.

The protection circuit AND gate 284 is also coupled to an under voltagelockout circuit 285 that compares the input voltage V_(in) to a limitingthreshold voltage and, when the input voltage V_(in) is less than thethreshold voltage, another signal is provided to the protection circuitAND gate 284. Thus, the output of the protection circuit AND gate 284provides an indication of either a high temperature condition or anunacceptably low input voltage V_(in) and disables the operation of thepower converter accordingly. For further protection during a faultcondition, when the main switch Q_(mn) is transitioned to anon-conducting state, the auxiliary switch Q_(aux) is enabled to conductby the action of the level shifting gate driver 220, thereby dischargingthe output capacitor C_(out) and providing further protection for theoutput voltage V_(out) of the power converter. As an example, the undervoltage lockout circuit 285 disables the operation of the powerconverter when the input voltage V_(in) is less than 2.5 volts. When theinput voltage V_(in) is less than about 2.6 volts, the linear regulator235 saturates “full on” and may lose its regulation capability, causinga drop in the bias voltage V_(bias). A sufficient voltage compliance,however, can be designed into the various devices in the power converterto enable proper operation when the bias voltage V_(bias) is slightlyless than the desired regulated value.

An output of the protection circuit AND gate 284 is also coupled to asoft start switch Q_(ss) through an inverter 287. The purpose of thesoft start switch Q_(ss) is to discharge the soft start capacitor Q_(ss)whenever a temperature within the power converter exceeds a limit or theinput voltage V_(in) is below a safe operating point for the power train230. Discharging the soft start capacitor C_(ss) essentially sets theset point for the output voltage V_(out) for the power train 230 tozero. The soft start capacitor C_(ss) may also be discharged by acircuit external to the power converter, such as by an external switch,to disable an operation of the power converter based on external systemrequirements.

The protection circuit AND gate 284 provides an input to the soft startAND gate 280, which is also coupled to the soft start comparator 275.The soft start AND gate 280 monitors an output of the soft startcomparator 275, which is coupled to the soft start capacitor C_(ss). Thesoft start AND gate 280 is coupled to a PWM AND gate 253 and configuredto disable the power train 230 whenever a voltage across the soft startcapacitor C_(ss) is less than a threshold value. In the presentembodiment, the threshold value, coupled to inverting input of the softstart comparator 275, is preferably about 0.8 volts. The threshold valuemay be derived from the bandgap reference circuit 260. Thus, a softstart circuit of the controller 210 includes, among other things, thesoft start capacitor C_(ss), the soft start switch Q_(ss) and the softstart comparator 275.

The controller 210 also includes other protective circuits such as anover current protection circuit 290. A sense switch Qs is coupled inparallel with the main switch Q_(mn) of the power train 230 and iscontrolled to conduct synchronously with the main switch Q. A senseresistor R_(s) is coupled in series with the sense switch Q_(s). Thus, acurrent that flows through the sense resistor R_(s) is a fraction of thecurrent flowing through the main switch Q_(mn) when the main switchQ_(mn) conducts. A voltage proportional to the sensed current isamplified by an operational amplifier (not shown) in the over currentprotection circuit 290 and compared to a threshold value. If thethreshold value of the current through the sense resistor R_(s) isexceeded, a disable signal is provided to the PWM AND gate 253. Thus,the over current protection circuit 290 can disable the operation of thepower train 230 whenever current through the main switch Q_(mn), whichalso generally flows through the output inductor L_(out), exceeds athreshold value.

A power good monitoring circuit 292 is coupled to the output of the softstart AND gate 280, and preferably provides a signal to the power goodconnection PWRGD of the power converter to provide an externalindication that the power converter is operating normally. In addition,the power good monitoring circuit 292 is also coupled to the referencevoltage V_(ref) from the reference voltage selector 265. When the outputof the reference voltage selector 265 is above a predetermined referencevoltage level and the output of the soft start AND gate 280 is high, theoutput of the power good monitoring circuit 292 is high to indicate anormal operation of the power converter. It should be understood thatcircuits that monitor internal voltages and the outputs of logic gatesare generally well known in the art. It should further be understoodthat circuits that monitor the operation of a power converter can beoptionally coupled to various operating points within the controller 210and the power train 230 of the power converter.

Thus, a power converter embodied in, or portions thereof, an integratedcircuit has been illustrated and described with respect to FIG. 2. Asdescribed above, the devices of the power converter may be constructedwith low voltage devices and higher voltage devices integrable in asemiconductor device using fine line processing. Thus, for reasons asstated below, not only can control and signal processing devices, buthigher voltage devices such as the switches of the driver and powertrain, can be integrated into a semiconductor device thereby furtherfacilitating the power converter incorporated into an integratedcircuit.

Turning now to FIG. 3, illustrated is a schematic diagram of anembodiment of a driver of a power converter embodied in, or portionsthereof, an integrated circuit constructed according to the principlesof the present invention. The driver is adapted to provide a drivesignal S_(DRV) to control a switch having a control voltage limit. Morespecifically and in the illustrated embodiment, the driver is a gatedriver that provides a gate drive signal S_(DRV) to, for instance, aP-channel MOSFET that exhibits a gate voltage limit (i.e., agate-to-source voltage limit) of 2.5 volts. The gate driver receives asignal (e.g., a pulse width modulated signal S_(PWM)) from a controller(see, for instance, the controller 120 illustrated and described withrespect to FIG. 1) and a complement of the signal (e.g., a complementarypulse width modulated signal S_(1-PWM)) from the controller.Additionally, the gate driver may provide a first gate drive signal anda second gate drive signal to drive multiple switches such as the mainand auxiliary switches Q_(mn), Q_(aux) of a power converter as describedabove. For purposes of the following discussion, however, the driverwill be described and is adapted to provide a gate drive signal S_(DRV).

The gate driver includes switching circuitry formed by a plurality ofdriver switches such as first, second, third, fourth, fifth, sixth,seventh and eighth driver switches Q_(DR1), Q_(DR2), Q_(DR3), Q_(DR4),Q_(DR5), Q_(DR6), Q_(DR7), Q_(DR8) coupled to a source of electricalpower for the power converter and the controller of the power converter.The gate driver is also coupled to a first bias voltage source thatprovides a first bias voltage V_(bias1), which may be internally orexternally generated and may depend on an input voltage V_(in) of thepower converter. For purposes of the discussion herein, it is assumedthat the first, second, third, fourth, fifth, sixth, seventh and eighthdriver switches Q_(DR1), Q_(DR2), Q_(DR3), Q_(DR4), Q_(DR5), Q_(DR6),Q_(DR7), Q_(DR8) have a low gate voltage limit and a higher voltagedrain. Thus, the first, second, third, fourth, fifth, sixth, seventh andeighth driver switches Q_(DR1), Q_(DR2), Q_(DR3), Q_(DR4), Q_(DRS),Q_(DR6), Q_(DR7), Q_(DR8) may exhibit a low gate voltage limit (e.g. 2.5volts) and at the same time handle drain-to-source voltages above thegate voltage limit thereof (e.g., ten volts).

To simplify the discussion, it is also assumed that the first, second,third, fourth, fifth, sixth, seventh and eighth driver switches Q_(DR1),Q_(DR2), Q_(DR3), Q_(DR4), Q_(DR5), Q_(DR6), Q_(DR7), Q_(DR8) exhibit agate threshold voltage of about is 0.5 volts, which is consistent with anumber of fine feature size, low voltage MOSFET designs. The gatethreshold voltage provides a voltage level above or below which(depending on the type) the first, second, third, fourth, fifth, sixth,seventh and eighth driver switches Q_(DR1), Q_(DR2), Q_(DR3), Q_(DR4),Q_(DR5), Q_(DR6), Q_(DR7), Q_(DR8) are enabled to conduct.

In the illustrated embodiment, the first, second, seventh and eighthdriver switches Q_(DR1), Q_(DR2), Q_(DR7), Q_(DR8) are N-channel MOSFETsand the third, fourth, fifth and sixth driver switches Q_(DR3), Q_(DR4),Q_(DR5), Q_(DR6) are P-channel MOSFETs. The drain terminals of thesecond, third and fifth driver switches Q_(DR2), Q_(DR3), Q_(DR5) arecoupled together at a first node n₁. The drain terminals of the first,fourth and sixth driver switches Q_(DR1), Q_(DR4), Q_(DR6) are coupledtogether at a second node n₂. While each of the first, second, seventhand eighth driver switches Q_(DR1), Q_(DR2), Q_(DR7), Q_(DR8) areillustrated with gate, source and drain terminals, it is also common foreach of the first, second, seventh and eighth driver switches Q_(DR1),Q_(DR2), Q_(DR7), Q_(DR8) to include a body terminal.

The gate driver is coupled between an input voltage V_(in) (e.g., anunregulated input voltage at a nominal five volts) of the powerconverter and ground, with a potential difference therebetween for thepurposes of this discussion of five volts. The source terminal of thethird and sixth driver switches Q_(DR3), Q_(DR6) are coupled to theinput voltage V_(in). The first bias voltage V_(bias1), assumed for thisdiscussion to be 2.5 volts with respect to the ground, is coupled to thegate terminal of the fourth and fifth driver switches Q_(DR4), Q_(DR5),and a return connection of the first bias voltage source is coupled tothe ground. The first bias voltage source may or may not be derived fromthe source of electrical power that provides the input voltage V_(in),depending on the application for the gate driver.

As illustrated, the seventh and eighth driver switches Q_(DR7), Q_(DR8)are parallel coupled to the fourth and fifth driver switches Q_(DR4),Q_(DR5), respectively. The seventh and eighth driver switches Q_(DR7),Q_(DR8) include a higher voltage source and a higher voltage drain andtypically exhibit a higher source-to-gate voltage handling capability(e.g., five volts) when the source is more positive than the gate and atthe same time handle drain-to-source voltages above the low gate voltagelimit thereof. The gate terminal of the seventh and eighth driverswitches Q_(DR7), Q_(DR8) are coupled together and to a second voltagebias source that provides a second bias voltage V_(bias2), which may beinternally or externally generated and may depend on an input voltageV_(in) of the power converter.

The gate driver, in the illustrated embodiment, can operate in a coupleof different modes of operation. For instance, when the input voltageV_(in) to the power converter is greater than an upper gate voltagelimit for a main switch Q_(mn) such as a P-channel MOSFET (see, as anexample, the power train of the power converter illustrated anddescribed with respect to FIG. 1) driven by the gate driver, thenvoltage protective features of the gate driver are enabled.

More specifically, when the pulse width modulated signal S_(PWM)provided to the second driver switch Q_(DR2) is high (i.e., when thepulse width modulated signal S_(PWM) is more positive than the gatethreshold voltage of 0.5 volts), the first node n₁ that couples thedrain terminals of the second and third driver switches Q_(DR2), Q_(DR3)is pulled low by the second driver switch Q_(DR2). The drain terminal ofthe fifth driver switch Q_(DR5) is also coupled to the first node n₁ andthe gate terminal thereof is coupled to the first bias voltage source.Thus, the source of the fifth driver switch Q_(DR5) is pulled down tothree volts (i.e., one gate threshold voltage value more positive thanthe first bias voltage V_(bias1)). The gate drive signal S_(DRV) istherefore pulled down two volts below the input voltage V_(in), which isa sufficient voltage to enable a switch such as the main switch Q_(mn),a P-channel MOSFET, illustrated and described with respect to the powertrain of the power converter of FIG. 1 to conduct.

When the complementary pulse width modulated signal S_(1-PWM) providedto the first driver switch Q_(DR1) is more positive than the gatethreshold voltage, the first driver switch Q_(DR1) is enabled to conductand the second node n₂ is pulled down to substantially the groundvoltage by an on-resistance of the first driver switch Q_(DR1). The gateof the third driver switch Q_(DR3) is pulled down to about three volts(i.e., one gate threshold voltage value more positive than the firstbias voltage V_(bias1)). Thus, the third driver switch Q_(DR3) isenabled to conduct and the drain thereof, coupled to first node n₁, ispulled up substantially to the input voltage V_(in). The fifth driverswitch Q_(DR5) is now enabled to conduct because the gate voltage ismore than one gate threshold voltage more negative than the drainthereof, and the source of the fifth driver switch Q_(DR5) is pulled upsubstantially to the input voltage V_(in). Therefore, the gate drivesignal S_(DRV) from the gate driver is also pulled up to substantiallythe input voltage V_(in), which is a sufficient voltage to transition aswitch such as the main switch Q_(mn), a P-channel MOSFET, illustratedand described with respect to the power train of the power converter ofFIG. 1 to a non-conducting state.

Accordingly, a type of level shifting gate driver has been introducedwith an improved level-shifting capability that can controllably raisethe gate voltage of an exemplary switch (e.g., a P-channel MOSFET) tosubstantially the input voltage V_(in) to transition the switch to anon-conducting state, and controllably reduce the gate voltage to alower voltage to enable the switch to conduct. Inasmuch as the gateterminal of the fifth driver switch Q_(DR5) is coupled to the first biasvoltage source, the fifth driver switch Q_(DR5) is transitioned to anon-conducting state when a voltage present on its source is less thanthe first bias voltage V_(bias1) plus its gate threshold voltage(treating the gate threshold voltage of a P-channel MOSFET as a positivenumber). If the gate driver properly applies the first bias voltageV_(bias1) (e.g., if the first bias voltage V_(bias1) is the inputvoltage V_(in) minus 2.5 volts and adjusted for the gate thresholdvoltage of the fifth driver switch Q_(DR5)), the gate drive signalS_(DRV) will not decrease more than 2.5 volts below input voltage V_(in)thereby not exceeding the gate voltage limit of the switch to be driven.The first bias voltage V_(bias1), therefore, is preferably dependent onthe input voltage V_(in). The gate terminal of the switch (again, aP-channel MOSFET) coupled to the gate driver will thus be protected bythe gate driver and, in particular, by the fifth driver switch Q_(DR5),which operatively provides a protective voltage limiting function.Finally, the gate driver is symmetrical and as the pulse width modulatedsignal S_(PWM) and complementary pulse width modulated signal S_(1-PWM)alternate, the conduction states and voltages within the gate driveralternate accordingly.

Additionally, in this mode of operation, the second bias voltageV_(bias2) provided to the gate terminals of the seventh and eighthdriver switches Q_(DR7), Q_(DR8) is at a ground potential. Since thesource terminals of the seventh and eighth driver switches Q_(DR7),Q_(DR8) are not coupled to a potential at or below the ground potential,the seventh and eighth driver switches Q_(DR7), Q_(DR8) are not enabledto conduct as a consequence of the grounded gate terminals thereof.Thus, under the aforementioned circumstances, the seventh and eighthdriver switches Q_(DR7), Q_(DR8) have little effect on the operation ofthe gate driver.

In another operating mode for the gate driver (enabled by the seventhand eighth driver switches Q_(DR7), Q_(DR8)), the input voltage V_(in)to the power converter is not greater than an upper gate voltage limitfor a main switch Q_(mn) such as a P-channel MOSFET (see, as an example,the power train of the power converter illustrated and described withrespect to FIG. 1) driven by the gate driver, then voltage protectivefeatures of the gate driver are not necessary. In this mode ofoperation, the clamping operation of the fifth driver switch Q_(DR5) onthe gate drive signal S_(DRV) is inoperative. More specifically, thegate terminal of the seventh and eighth driver switches Q_(DR7), Q_(DR8)are coupled to a suitably high potential such as the input voltageV_(in). As a result, the seventh and eighth driver switches Q_(DR7),Q_(DR8) are enabled to conduct. Thus, the gate drive signal S_(DRV) iscoupled to ground potential by an on resistance of the second and eighthdriver switches Q_(DR2), Q_(DR8) when the main switch Q_(mn), aP-channel MOSFET as discussed above, driven by the gate driver isenabled to conduct. The gate driver, therefore, selectively providesadditional flexibility by altering a voltage applied to an inputthereof, consequently accommodating an input voltage V_(in) above orbelow a gate voltage limit of a switch driven therefrom. Additionally,for a more detailed analysis of this embodiment of the driver, see U.S.patent application Ser. No. 10/767,540, entitled “Driver for a PowerConverter and Method of Driving a Switch Thereof,” to Dwarakanath, etal., which is incorporated herein by reference.

Turning now to FIG. 4, illustrated is a cross sectional view of anembodiment of a semiconductor device employable in an integrated circuitconstructed according to the principles of the present invention. Asemiconductor substrate (also referred to as a “substrate”) 415 of thesemiconductor device is divided into four dielectrically separated areasto accommodate, in the illustrated embodiment, four transistors (e.g.,MOSFETs) located thereon. More specifically, the substrate 415 canaccommodate a PMOS device and a NMOS device that operate as low voltagedevices within, for instance, a controller of a power converter (i.e.,the control and signal processing devices). Additionally, the substrate415 can accommodate a P-LDMOS device and a N-LDMOS device that operateas higher voltage devices within, for instance, a power train and driverof a power converter (i.e., the power switches and driver switches).

The semiconductor device also includes shallow trench isolation regions410 formed within the substrate 415 to provide dielectric separationbetween the devices implemented on the substrate 415. The shallow trenchisolation regions 410 are formed by masking the substrate 415 and usinga photoresist to define the respective regions therein. The shallowtrench isolation regions 410 are then etched and backfilled with adielectric such as silicon dioxide, silicon nitride, a combinationthereof, or any other suitable dielectric material. Then, the substrate415 and the shallow trench isolation regions 410 are planarized by alapping process.

A buried layer (e.g., a N-type buried layer) 420 is recessed within thesubstrate 415 in the area that accommodates the P-LDMOS device and theN-LDMOS device. The N-type buried layer 420 is formed by a deep ionimplantation process (e.g., at a controlled voltage of about 200kiloelectronvolts) of an appropriate dopant specie such as arsenic orphosphorus and results in a doping concentration profile, preferably ina range of 1×10⁸ to 1×10²⁰ atoms/cm³. The N-type buried layer 420 ispreferably located approximately one micrometer below a top surface ofthe substrate 415, and is annealed (e.g., at 600 to 1200 degreesCelsius) as necessary to provide the proper distribution of theimplanted ion specie.

The semiconductor device also includes wells (e.g., N-type wells) 425formed in the substrate 415 in the areas that accommodate the PMOSdevice and the P-LDMOS device, and under the shallow trench isolationregions 410 above the N-type buried layer 420 (for the P-LDMOS device).The N-type wells 425 are formed to provide electrical isolation for thePMOS device and the P-LDMOS device and operate cooperatively with theN-type buried layer 420 (in the case of the P-LDMOS device) and theshallow trench isolation regions 410 to provide the isolation. Asillustrated, the N-type well 425 above the N-type buried layer 420 doesnot cover the entire area that accommodates the P-LDMOS device in thesubstrate 415 between the shallow trench isolation regions 410 thereof.A photoresist mask defines the lateral areas for ion implantationprocess to form the N-type wells 425. After the ion implantationprocess, the implanted specie is diffused by annealing the substrate 415at elevated temperature. An appropriate dopant specie such as arsenic orphosphorus can be used to form the N-type wells 425, preferably, butwithout limitation, in a retrograde doping concentration profile withapproximately 1×10¹⁷ atoms/cm³ in the middle, and a higher dopingconcentration profile at the surface as well as at the bottom. Theadvantages of forming the N-type well 425 in the substrate 415 within aportion of the area that accommodates the P-LDMOS device will becomemore apparent for the reasons as set forth below.

The semiconductor device includes additional wells (e.g., P-type wells)430 formed in the substrate 415 between the shallow trench isolationregions 410 substantially in the areas that accommodate the NMOS deviceand N-LDMOS device. While the P-type well 430 above the N-type buriedlayer 420 covers the entire area that accommodates the N-LDMOS device inthe substrate 415 between the shallow trench isolation regions 410thereof, it is well within the broad scope of the present invention todefine the P-type well 430 to cover a portion of the area thataccommodates the N-LDMOS device in the substrate 415.

A photoresist mask defines the lateral areas for the ion implantationprocess to form the P-type wells 430. After the ion implantationprocess, the implanted specie is diffused by an annealing the substrate415 at an elevated temperature. An appropriate dopant specie such asboron can be used to form the P-type wells 430, preferably resulting ina retrograde doping concentration profile with approximately 1×10¹⁷atoms/cm³ in the middle, and a higher doping concentration profile atthe top surface as well as at the bottom. Analogous to the N-type wells425, a width of the P-type wells 430 may vary depending on theparticular devices and application and, as those skilled in the artknow, may be laterally defined by the photoresist mask. For instance,while the P-type well 430 above the N-type buried layer 420 covers theentire area that accommodates the N-LDMOS device in the substrate 415between the shallow trench isolation regions 410 thereof, it is wellwithin the broad scope of the present invention to define the P-typewell 430 to cover a portion of the area that accommodates the N-LDMOSdevice in the substrate 415.

The semiconductor device also includes gates 440 for the PMOS, NMOS,P-LDMOS and N-LDMOS devices located over a gate dielectric layer 435 andincluding gate sidewall spacers 455 about the gates 440 thereof. Thedielectric material for the gate dielectric layer 435 is typicallysilicon dioxide with a thickness of about five nanometers for devicesemploying about 0.25 micrometer feature sizes and operating at low gatevoltages (e.g., 2.5 volts). Assuming the gate-to-source voltage limit ofthe P-LDMOS and N-LDMOS devices is limited to a lower voltage (e.g., 2.5volts) and the PMOS and NMOS devices operate at the same voltage, thenthe gate dielectric layer 435 can be formed with dimensions as set forthabove. Preferably, the gate dielectric layer 435 is constructed with auniform thickness to provide a gate-to-source voltage rating for thedevices of approximately 2.5 volts that completely or nearly completelysaturates the forward conduction properties of the device. Of course,the aforementioned voltage range for the devices is provided forillustrative purposes only and other voltage ranges are within the broadscope of the present invention.

A polysilicon layer is deposited over a surface of the gate dielectriclayer 435 and doped N-type or P-type, using an appropriate dopingspecie. The polysilicon layer is annealed at an elevated temperature toproperly diffuse the dopant. A photoresist mask is employed with an etchto define the lateral dimensions to define the gates 440. The thicknessof the gates 440 may range from about 100 to about 500 nanometers, butmay be even smaller or larger depending on the application. The gatesidewall spacers 455, which may be formed from an oxide or otherdielectric material, are generally formed by depositing a nitridefollowed by an etching process.

The N-LDMOS device includes lightly doped regions (e.g., N-type lightlydoped regions) 445 for the source and the drain thereof. The P-LDMOSdevice also includes lightly doped regions (e.g., P-type lightly dopedregions) 450 for the source and the drain thereof. In the presentembodiment, the N-type and P-type lightly doped regions 445, 450 providehigher voltage sources and drains for the N-LDMOS and P-LDMOS devices,respectively. As a result, not only can the N-LDMOS and P-LDMOS deviceshandle higher voltages from the drain-to-source thereof, but the devicescan handle a higher voltage from a source-to-gate thereof when thesource is more positive than the gate 440. It is recognized that thewidth of the N-type and P-type lightly doped regions 445, 450 may beindividually varied to alter the breakdown voltage characteristics ofthe respective N-LDMOS and P-LDMOS devices without departing from thescope of the present invention.

The N-type and P-type lightly doped drain regions 445, 450 may be formedemploying an ion implantation process in connection with a photoresistmask to define the lateral dimensions thereof. Additionally, anannealing process at elevated temperatures distributes the implanted ionspecie. The N-type and P-type lightly doped drain regions 445,450 arepreferably doped, without limitation, to about 1×10¹⁶ to 1×10¹⁷atoms/cm³.

The semiconductor device also includes heavily doped regions (e.g.,N-type heavily doped regions) 460 for the source and drain of the NMOSdevice that preferably have a different doping concentration profilethan heavily doped regions (e.g., N-type heavily doped regions) 462 forthe source and drain of the N-LDMOS device. The N-type heavily dopedregions 460 for the NMOS device are formed within the P-type well 430thereof and form the source and the drain for the NMOS device.Additionally, the N-type heavily doped regions 462 for the N-LDMOSdevice are formed within the P-type well 430 thereof and form a portionof the source and the drain for the N-LDMOS device.

The N-type heavily doped regions 460, 462 may be advantageously formedwith an ion implantation process using dopant specie such as arsenic orphosphorus. The doping process includes a photoresist mask to definelateral dimensions of the N-type heavily doped regions 460, 462 and anannealing process at elevated temperature to properly distribute theimplanted species. The N-type heavily doped region 460 for the sourceand drain of the NMOS device is doped, without limitation, to be greaterthan about 1×10¹⁹ atoms/cm³. The N-type heavily doped region 462 for thesource and drain of the N-LDMOS device is doped, without limitation, tobe greater than about 5×10¹⁹ atoms/cm³.

The semiconductor device also includes heavily doped regions (e.g.,P-type heavily doped regions) 465 for the source and drain of the PMOSdevice that preferably have a different doping concentration profilethan heavily doped regions (e.g., P-type heavily doped regions) 467 forthe source and drain of the P-LDMOS device. The P-type heavily dopedregions 465 for the PMOS device are formed within the N-type well 425thereof and form the source and the drain for the PMOS device.Additionally, the P-type heavily doped regions 467 for the P-LDMOSdevice are formed within the N-type well 425 or in regions adjacent theN-type well 425 thereof and form a portion of the source and the drainfor the P-LDMOS device.

The P-type heavily doped regions 465, 467 may be advantageously formedwith an ion implantation process using dopant specie such as boron. Thedoping process includes a photoresist mask to define lateral dimensionsof the P-type heavily doped regions 465, 467 and an annealing process atelevated temperature to properly distribute the implanted species. TheP-type heavily doped region 465 for the source and drain of the PMOSdevice is doped, without limitation, to be greater than about 1×10¹⁹atoms/cm³. The P-type heavily doped region 467 for the source and drainof the P-LDMOS device is doped, without limitation, to be greater thanabout 5×10¹⁹ atoms/cm³.

In the illustrated embodiment, the N-type well 425 above the N-typeburied layer 420 does not cover the entire area that accommodates theP-LDMOS device in the substrate 415 between the shallow trench isolationregions 410 thereof. In particular, the N-type well 425 is located underand within a channel region 470, and the N-type well 425 and N-typeburied layer 420 are oppositely doped in comparison to the P-typelightly and heavily doped regions 450, 467. Thus, doped regions (e.g., aP-type doped regions; also generally referred to as a “doped region andanother doped region”) 472, 474 extend between the P-type heavily dopedregions 467 and the N-type well 425 of the P-LDMOS device and have adoping concentration profile less than a doping concentration profile ofthe P-type heavily doped regions 467. While the P-type heavily dopedregions 467 preferably have the same doping concentration profiles, itis well within the broad scope of the present invention that the P-typeheavily doped region 467 for the source has a different dopingconcentration profile than the counterpart of the drain. The sameprinciple applies to other like regions of the devices of thesemiconductor device.

In the illustrated embodiment, the P-type doped regions 472, 474 happento be embodied in the substrate 415 which has a doping concentrationprofile between 1×10⁴ and 1×10¹⁶ atoms/cm³. Employing the substrate 415as the P-type doped regions 472, 474 provides an opportunity to omit amasking and a processing step in the manufacture of the semiconductordevice. In yet another alternative embodiment, the P-type doped regions472, 474 may be formed by an ion implantation process prior toimplanting the P-type heavily doped regions 467 for the source and thedrain of the P-LDMOS device. Of course, the P-type doped regions 472,474 may be formed with any doping concentration profile less than theP-type heavily doped regions 467.

Incorporating the P-type doped regions 472, 474 into the P-LDMOS devicefurther increases a breakdown voltage between the P-type heavily dopedregions 467 and the N-type well 425 of the P-LDMOS device. The P-LDMOSdevice, therefore, exhibits a higher drain-to-source voltage handingcapability due to the higher breakdown voltage thereof and provides ahigher source-to-gate voltage handling capability when the source ismore positive than the gate 440. It should be understood that while thedoped regions have been described with respect to the P-LDMOS device,the principles are equally applicable to the N-LDMOS device and, forthat matter, other transistors of analogous construction.

Additionally, the P-LDMOS and N-LDMOS devices illustrated and describedwith respect to FIG. 4 are referred to as symmetrical devices. In otherwords, the symmetrical nature of the source and drain of thesemiconductor device of FIG. 4 provide for a symmetrical device. Ofcourse, those skilled in the art should understand that the dimensionsof the source and drain (including the lightly and heavily dope regionsthereof) may vary and still fall within the broad scope of the presentinvention. The semiconductor device also includes metal contacts 485defined by dielectric regions 480 formed over salicide layers (one ofwhich is designated 475) for the gate, source and drain of the PMOS,NMOS, P-LDMOS and N-LDMOS devices.

The development of a semiconductor device as described herein retainsthe fine line structures and accommodates an operation at highervoltages and with higher switching frequencies (e.g., five megahertz).By introducing a doped region(s) between the heavily doped region andoppositely doped well, the LDMOS device exhibits a high voltage handlingcapability from the drain to the source thereof (e.g., ten volts). Atthe same time, the higher voltage device is constructed employing alimited number of additional processing steps. Moreover, the LDMOSdevice may exhibit a low level gate-to-source voltage limit (e.g., 2.5volts) and at the same time handle drain-to-source voltages above thegate-to-source voltage limit thereof. Alternatively, the LDMOS devicemay exhibit a higher level source-to-gate voltage handling capability(e.g., five volts) when the source is more positive than the gate and atthe same time handle drain-to-source voltages above the low levelgate-to-source voltage limit thereof. In other words, the LDMOS devicecan switch the larger currents normally associated with a power train ofa power converter by appropriately designing selected regions thereof asset forth above. For a better understanding of the semiconductor deviceillustrated and described with respect to FIG. 4, see U.S. patentapplication Ser. No. 10/767,684, entitled “Laterally Diffused MetalOxide Semiconductor Device and Method of Forming the Same,” to Lotfi, etal., which is incorporated herein by reference.

Turning now to FIGS. 5A to 5H, illustrated are cross sectional views ofan embodiment of constructing a micromagnetic device employable in anintegrated circuit constructed according to the principles of thepresent invention. According to one embodiment, a planar inductor,optionally a double-spiral planar inductor, is formed as an example of amicromagnetic device. As illustrated in FIG. 5A, a semiconductorsubstrate (also referred to as a “substrate”) 500 is provided with apassivation or substrate insulation layer 505 formed thereon, typicallyan oxide having a thickness of about two micrometers. Additionally, asemiconductor device 510 (such as the semiconductor device illustratedand described with respect to FIG. 4) is formed within the substrate500. Additionally, a plurality of vias (one of which is designated 515)are formed within the substrate insulation layer 505 to provideelectrical connectivity between the semiconductor device 510 and themicromagnetic device, and to provide a ground connection for themagnetic cores thereof. Of course, in keeping with the spirit of thepresent invention, the integrated circuit employing the micromagneticdevice may include a controller, driver and power train embodied in, orportions thereof, the integrated circuit.

Turning now to FIG. 5B, an insulation and planarization layer isembodied in a first photoresist layer 520 formed over the substrateinsulation layer 505 and the substrate 500. Typically, the firstphotoresist layer 520 is patterned to maintain the integrity of the vias515. Advantageously, the first photoresist layer 520 is a Novolac-typepositive photoresist layer sensitive to ultraviolet light such as aAZ-4000 series sold by AZ Electronic Materials, a division of ClariantCorporation of Charlotte, N.C. The first photoresist layer 520 issubstantially inert to aggressive chemical environments such aselectroplating solutions. Good adhesion to a variety of metals is alsodesirable, since metal seed layers for electroplating are often formedon the first photoresist layer 520.

In addition, when it is desirable to incorporate the first photoresistlayer 520 as an insulation and planarization layer, the firstphotoresist layer 520 advantageously exhibits desirable mechanical andelectrical properties. More specifically, in multilayer components, thefirst photoresist layer 520 advantageously has the ability to provide aplanar surface for subsequent lithography and layer formation. Also, thedesirable electrical properties include an acceptable dielectricconstant when cured. Depending on the desired thickness of the firstphotoresist layer 520, it is possible that the first photoresist layer520 constitutes several thin layers. Patterning of the first photoresistlayer 520 is performed according to conventional techniques. Curing isused to render the first photoresist layer 520 substantially inert tosubsequent environments such as electroplating solutions.

Turning now to FIG. 5C, after the first photoresist layer 520 ispatterned, a first metallic seed layer 525 is formed to act as a seedlayer for subsequent electroplating of a lower magnetic core of themicromagnetic device. An exemplary material for the first metallic seedlayer 525 is a two layer titanium/gold film, e.g., about 125 to 300angstroms of titanium followed by 500 to 3000 angstroms of gold. Thegold exhibits desirable resistivity and chemical properties, and thetitanium enhances adhesion of the gold to the cured first photoresistlayer 520. The gold and titanium are typically deposited by sputteringor electron beam deposition. It is also possible to use atitanium/copper film as the first metallic seed layer 525.

Turning now to FIG. 5D, a patterned photoresist layer 530 is formed andpatterned for subsequent formation of the lower magnetic core. Thethickness of the patterned photoresist layer 530 is based on the desiredthickness of the magnetic core, i.e., the magnetic material is generallyelectroplated up to the top surface of the patterned photoresist layer530.

Turning now to FIG. 5E, after formation of the patterned photoresistlayer 530, a lower magnetic core 535 is formed by electroplating on tothe first metallic seed layer 525, and the patterned photoresist layer530 is then removed. The lower magnetic core 535 is formed from anysuitable magnetic material, typically a soft magnetic material for aninductor and transformer applications (e.g., Permalloy). Advantageously,the lower magnetic core 535 is formed from a iron alloy. The propertiesof a desirable magnetic core material include relatively low coercivity,relatively high electrical resistivity, and relatively high saturationmagnetization.

Turning now to FIG. 5F, a second photoresist layer 540 is formed asanother insulation and planarization layer and is patterned to retainthe integrity of the vias 515. Thereafter, a second metallic seed layer545, typically of the same material and properties as described above,is formed on the second photoresist layer 540. In particular, the secondmetallic seed layer 545 is patterned in a manner that avoids creation ofa short circuit between spirals of the to-be-formed conductive coil orwinding. Alternatively, portions of the first and second metallic seedlayers 525, 545 that may create such short circuits may be removedduring a later step.

Turning now to FIG. 5G, a third photoresist layer 550 is formed toprovide a mold for formation of a conductive coil 560, which istypically a copper coil. The conductive coil 560 is formed byelectroplating on to the second metallic seed layer 545. A thickness ofthe conductive coil 560 and the third photoresist layer 550 is typicallyabout 30 micrometers. The third photoresist layer 550 is not removed,but instead remains to act as an electrical insulator and a planar,mechanical support layer within the micromagnetic device.

Turning now to FIG. 5H, a fourth photoresist layer 570 is formed toprovide an another insulation and planarization layer. It should beunderstood that it is possible to cure both the third and fourthphotoresist layers 550, 570 at the same time. A third metallic seedlayer 580 is formed for electroplating and formed from materialsanalogous to the first metallic seed layer 525 described above. Anotherpatterned photoresist layer (not shown) is then formed to provide a moldfor the subsequent electroplating of an upper magnetic core 590 andhaving a thickness selected to provide the upper magnetic core 590 ofsubstantially the same thickness. Finally, the upper magnetic core 590is electroplated with a material and technique as described above withrespect to the lower magnetic core 535. The another photoresist layer isthen removed. Optionally, an additional insulation layer may be formedover the upper magnetic core 590 for protection purposes.

Turning now to FIG. 6, illustrated is an isometric view of an embodimentof a micromagnetic device employable in an integrated circuitconstructed according to the principles of the present invention. Themicromagnetic device is formed on a semiconductor substrate (alsoreferred to as a “substrate”) 610 with lower and upper magnetic cores620, 630, a spiral conductive coil 640 (a single spiral is shown), alongwith intermediate cured photoresist layers 650, 660, 670. In oneembodiment, the lower and upper magnetic cores 620, 630 are formed froman iron-cobalt-phosphorus alloy, and are about five micrometers thick.The spiral conductive coil or winding 640 is formed from copper in athickness of about 30 micrometers and the photoresist layers 650, 660,670 are formed from one or more AZP-4000 series photoresist layers asmentioned above.

Numerous variations of this particular embodiment are possible includingperformance of the steps above in different order and with additional oralternative layers, and should be apparent to those skilled in the art.Additionally, for a more detailed analysis of the micromagnetic deviceas described herein, see U.S. Pat. No. 6,495,019 entitled “DeviceComprising Micromagnetic Components for Power Applications and Processfor Forming Device,” to Filas, et al., issued Dec. 17, 2002 and “Issuesand Advances in High-Frequency Magnetics for Switching Power Supplies,”by Lotfi, et al., Proceedings of the IEEE, Vol. 89, No. 6, pp. 833-845,June 2001, both of which are incorporated herein by reference.

Turning now to FIG. 7, illustrated is a cross sectional view of anembodiment of an output filter employable in an integrated circuitconstructed according to the principles of the present invention. In theillustrated embodiment, the output filter includes a capacitor coupledto an inductor embodied in a micromagnetic device. The output filter isconstructed on a semiconductor substrate (also referred to as a“substrate,” and composed of, for instance, silicon, glass, ceramic orthe like) 710 having a passivation layer (e.g., silicon dioxide) 720formed thereon using conventional formation processes such as a thermalgrowing process.

The micromagnetic device includes a first and second conductive windinglayer (composed of, for instance, aluminum or any other conductivematerial) 740, 760 surrounded by first, second and third insulativelayers or insulators 730, 750, 770. The micromagnetic device alsoincludes a metallic layer 780 that provides an adequate bond between aferromagnetic core 790 and the insulators 730, 750, 770 coupled to thesubstrate 710 to facilitate the fabrication of the thereof. Themicromagnetic device still further includes a plurality of inner-layervias that provide multiple paths between layers of the micromagneticdevice and a terminal 796 for connection to another device.

The capacitor includes first and second capacitor plates 745, 755 and adielectric layer 735 located between the first and second capacitorplates 745, 755. The capacitor and micromagnetic device are electricallycoupled as illustrated by the conductive layers running therebetween.The capacitor also includes a plurality of inner-layer vias that providemultiple paths between the first and second plates 745, 755 of thecapacitor and a terminal 797 for connection to another device. Anembodiment of a micromagnetic device is disclosed in U.S. Pat. No.6,118,351 entitled “Micromagnetic Device for Power ProcessingApplications and Method of Manufacture Therefor,” to Kossives, et al.,issued Sep. 12, 2000, and several embodiments of filter circuits aredisclosed in U.S. Pat. No. 6,255,714 entitled “Integrated Circuit Havinga Micromagnetic Device Including a Ferromagnetic Core and Method ofManufacture Therefor,” to Kossives, et al., issued Jul. 3, 2001, both ofwhich are incorporated by reference.

Thus, a power converter embodied, or portions thereof, in an integratedcircuit and related methods of constructing the same with readilyattainable and quantifiable advantages has been introduced. Thoseskilled in the art should understand that the previously describedembodiments of the integrated circuit including the power converter andportions thereof embodied in the integrated circuit and related methodsof constructing the same are submitted for illustrative purposes only.In addition, other embodiments capable of producing an integratedcircuit employable with higher voltage devices and low voltage devicesintegrable within a semiconductor device are well within the broad scopeof the present invention. While the integrated circuit has beendescribed in the environment of a power converter, the integratedcircuit may also apply to other systems such as a power amplifier, motorcontroller, and a system to control an actuator in accordance with astepper motor or other electromechanical device.

For a better understanding of integrated circuits, semiconductor devicesand methods of manufacture therefor see “Semiconductor DeviceFundamentals,” by R. F. Pierret, Addison-Wesley (1996); “Handbook ofSputter Deposition Technology,” by K. Wasa and S. Hayakawa, NoyesPublications (1992); “Thin Film Technology,” by R. W. Berry, P. M. Halland M. T. Harris, Van Nostrand (1968); “Thin Film Processes,” by J.Vossen and W. Kern, Academic (1978); and “Handbook of Thin FilmTechnology,” by L. Maissel and R. Glang, McGraw Hill (1970). For abetter understanding of power converters, see “Modem DC-to-DC SwitchmodePower Converter Circuits,” by Rudolph P. Severns and Gordon Bloom, VanNostrand Reinhold Company, New York, N.Y. (1985) and “Principles ofPower Electronics,” by J. G. Kassakian, M. F. Schlecht and G. C.Verghese, Addison-Wesley (1991). The aforementioned references areincorporated herein by reference in their entirety.

Also, although the present invention and its advantages have beendescribed in detail, it should be understood that various changes,substitutions and alterations can be made herein without departing fromthe spirit and scope of the invention as defined by the appended claims.For example, many of the processes discussed above can be implemented indifferent methodologies and replaced by other processes, or acombination thereof.

Moreover, the scope of the present application is not intended to belimited to the particular embodiments of the process, machine,manufacture, composition of matter, means, methods and steps describedin the specification. As one of ordinary skill in the art will readilyappreciate from the disclosure of the present invention, processes,machines, manufacture, compositions of matter, means, methods, or steps,presently existing or later to be developed, that perform substantiallythe same function or achieve substantially the same result as thecorresponding embodiments described herein may be utilized according tothe present invention. Accordingly, the appended claims are intended toinclude within their scope such processes, machines, manufacture,compositions of matter, means, methods, or steps.

1. An integrated circuit employable with a power converter, comprising:a power switch of a power train of said power converter formed on asemiconductor substrate; and a driver switch of a driver configured toprovide a drive signal to said power switch and embodied in atransistor, including: a gate located over a channel region recessedinto said semiconductor substrate, a source/drain including a lightlydoped region located adjacent said channel region and a heavily dopedregion located adjacent said lightly doped region, an oppositely dopedwell located under and within said channel region, and a doped region,located between said heavily doped region and said oppositely dopedwell, having a doping concentration profile less than a dopingconcentration profile of said heavily doped region.
 2. The integratedcircuit as recited in claim 1 wherein said power switch is a transistor,including: a gate located over a channel region recessed into saidsemiconductor substrate, a source/drain including a lightly doped regionlocated adjacent said channel region and a heavily doped region locatedadjacent said lightly doped region, and an oppositely doped well locatedunder and within said channel region.
 3. The integrated circuit asrecited in claim 2 wherein said transistor employable as said powerswitch further includes a doped region, located between said heavilydoped region and said oppositely doped well, having a dopingconcentration profile less than a doping concentration profile of saidheavily doped region thereof.
 4. The integrated circuit as recited inclaim 1 wherein said transistor, further includes: another source/drainincluding a lightly doped region located adjacent said channel regionand a heavily doped region located adjacent said lightly doped region,and another doped region, located between said heavily doped region ofsaid another source/drain and said oppositely doped well, having adoping concentration profile less than a doping concentration profile ofsaid heavily doped region of said another source/drain.
 5. Theintegrated circuit as recited in claim 1 wherein said transistor furtherincludes a gate dielectric layer underlying said gate and gate sidewallspacers about said gate, said transistor further including metalcontacts formed over a salicide layer on said gate and saidsource/drain.
 6. The integrated circuit as recited in claim 1 wherein anoppositely doped buried layer is located under said doped region.
 7. Theintegrated circuit as recited in claim 1 further comprising acomplementary metal oxide semiconductor device, formed on saidsemiconductor substrate, that includes a source/drain having a heavilydoped region with a doping concentration profile different from saiddoping concentration profile of said heavily doped region of saidsource/drain of said transistor.
 8. The integrated circuit as recited inclaim 1 further comprising a complementary metal oxide semiconductordevice employable in a controller of said power converter.
 9. Theintegrated circuit as recited in claim 1 further comprising a switch ina controller of said power converter formed by a transistor, including:a gate located over a channel region recessed into said semiconductorsubstrate, a source/drain including a lightly doped region locatedadjacent said channel region and a heavily doped region located adjacentsaid lightly doped region, and an oppositely doped well located underand within said channel region.
 10. The integrated circuit as recited inclaim 1 further comprising a micromagnetic device.
 11. The integratedcircuit as recited in claim 1 further comprising an output filter at anoutput of said power converter.
 12. The integrated circuit as recited inclaim 1 further comprising a capacitor associated with a controller ofsaid power converter.
 13. The integrated circuit as recited in claim 1further comprising a linear regulator within a controller of said powerconverter.
 14. The integrated circuit as recited in claim 1 furthercomprising a soft start circuit within a controller of said powerconverter.
 15. The integrated circuit as recited in claim 1 furthercomprising protection circuits within a controller of said powerconverter.
 16. An integrated circuit employable with a power converter,comprising: a power switch of a power train of said power converterformed on a semiconductor substrate; a complementary metal oxidesemiconductor device employable in a controller configured to provide asignal to control a duty cycle of said power switch and formed on saidsemiconductor substrate; and a driver switch of a driver configured toprovide a drive signal to said power switch as a function of said signalfrom said controller and embodied in a laterally diffused metal oxidesemiconductor device, including: a gate located over a channel regionrecessed into said semiconductor substrate, a source/drain including alightly doped region located adjacent said channel region and a heavilydoped region located adjacent said lightly doped region, an oppositelydoped well located under and within said channel region, and a dopedregion, located between said heavily doped region and said oppositelydoped well, having a doping concentration profile less than a dopingconcentration profile of said heavily doped region.
 17. The integratedcircuit as recited in claim 16 wherein said power switch is referencedto a voltage level different from said driver switch and said powerswitch is a transistor, including: a gate located over a channel regionrecessed into said semiconductor substrate, a source/drain including alightly doped region located adjacent said channel region and a heavilydoped region located adjacent said lightly doped region, and anoppositely doped well located under and within said channel region. 18.The integrated circuit as recited in claim 17 wherein said transistoremployable as said power switch further includes a doped region, locatedbetween said heavily doped region and said oppositely doped well, havinga doping concentration profile less than a doping concentration profileof said heavily doped region thereof.
 19. The integrated circuit asrecited in claim 16 wherein said laterally diffused metal oxidesemiconductor device, further includes: another source/drain including alightly doped region located adjacent said channel region and a heavilydoped region located adjacent said lightly doped region, and anotherdoped region, located between said heavily doped region of said anothersource/drain and said oppositely doped well, having a dopingconcentration profile less than a doping concentration profile of saidheavily doped region of said another source/drain.
 20. The integratedcircuit as recited in claim 16 wherein said laterally diffused metaloxide semiconductor device further includes a gate dielectric layerunderlying said gate and gate sidewall spacers about said gate, saidlaterally diffused metal oxide semiconductor device further includingmetal contacts formed over a salicide layer on said gate and saidsource/drain.
 21. The integrated circuit as recited in claim 16 whereinan oppositely doped buried layer is located under said doped region. 22.The integrated circuit as recited in claim 16 wherein said complementarymetal oxide semiconductor device includes a source/drain having aheavily doped region with a doping concentration profile different fromsaid doping concentration profile of said heavily doped region of saidsource/drain of said laterally diffused metal oxide semiconductordevice.
 23. The integrated circuit as recited in claim 16 wherein saidcontroller further includes a laterally diffused metal oxidesemiconductor device, including: a gate located over a channel regionrecessed into said semiconductor substrate, a source/drain including alightly doped region located adjacent said channel region and a heavilydoped region located adjacent said lightly doped region, and anoppositely doped well located under and within said channel region. 24.The integrated circuit as recited in claim 16 further comprising amicromagnetic device.
 25. The integrated circuit as recited in claim 16further comprising a shallow trench isolation region formed within saidsemiconductor substrate and proximate said doped region.
 26. Theintegrated circuit as recited in claim 16 further comprising an outputfilter at an output of said power converter.
 27. The integrated circuitas recited in claim 16 further comprising a capacitor associated withsaid controller of said power converter.
 28. The integrated circuit asrecited in claim 16 wherein said controller further includes a linearregulator.
 29. The integrated circuit as recited in claim 16 whereinsaid controller further includes a soft start circuit.
 30. Theintegrated circuit as recited in claim 16 wherein said controllerfurther includes protection circuits.